3 Basic Radar Systems – Military Avionics Systems

Basic Radar Systems

3.1 Basic Principles of Radar

The original concept of radar was demonstrated by laboratory experiments carried out by Heinrich Hertz in the 1880s. The term RADAR stands for Radio Aid to Detection And Ranging. Hertz demonstrated that radio waves had the same properties as light (apart from the difference in frequency). He also showed that the radio waves could be reflected from a metal object and could be refracted by a dielectric prism mimicking the behaviour of light.

The concept of radar was known and was being investigated in the 1930s by a number of nations, and the British introduced a ground-based early warning system called Chain Home. In the late 1930s, as part of the world’s first integrated air defence, this system has been credited with the winning of the Battle of Britain in 1940. The invention of the magnetron in 1940 gave the ability to produce power at higher frequencies and allowed radar to be adopted for airborne use. The first application was to airborne interception (AI) radars fitted to fighter aircraft to improve the air defence of Great Britain when used in conjunction with the Chain Home system. By the end of WWII, rudimentary ground-mapping radars had also been introduced under the dubious name of H2S. Echoes of War (Lovell, 1992) gives a fascinating account of the development of radar during the war. Since that time, radar has evolved to become the primary sensor on military aircraft and is widely used in civil aviation as a weather radar able to warn the flight crew of impending heavy precipitation or turbulence. For further information, see Pilot’s Handbook – Honeywell Radar RDR-4B.

Since that time, enormous advances have been made in airborne radars. Fighter aircraft carry multimode radars with advanced pulse Doppler (PD), track-while-scan (TWS) and synthetic aperture (SA) modes that impart an awesome capability. Larger aircraft with an airborne early warning (AEW), such as the E-3, carry large surveillance radars aloft with aerial dishes in excess of 20 ft in diameter. At the other end of the scale, attack helicopters such as the AH-64 C/D Longbow Apache deploy near-millimetric radars in a ‘doughnut’ on top of the rotor, measuring no more than 3 ft across (Figure 3.1).

Figure 3.1 Contrasting airborne radar applications. (I. Moir)

The performance and application of radar is highly dependent upon the frequency of operation. Figure 3.2 shows the range of electromagnetic (em) applications used in modern military avionics systems. The applications may be grouped into three categories in ascending order of frequency:

  1. Communications and Navaids, more correctly referred to as Communications, Navigation and Identification (CNI), operating in the band from 100 kHz to just over 1 GHz. CNI systems are addressed in Chapter 7.
  2. Airborne radar from ∼400 MHz to a little under 100 GHz. This is the subject of this chapter and of Chapter 4.
  3. Electrooptics (EO) including visible light in the band from a little over 10 000 GHz (10 THz) extending to just over 1 000 000 GHz (1000 THz). The frequency numbers are so high at this end of the spectrum that wavelength tends to be used instead. The EO band encompasses visible light, infrared (IR) and laser systems which are described in Chapter 5.

Focusing on the airborne radar systems that are the subject of this and the next chapter, these cover the frequency range from ∼400 MHz to 94 GHz, as shown in Figure 3.3. This illustrates some of the major areas of the spectrum as used by airborne platforms. In ascending order of frequency, typical applications are:

  • E-2C Hawkeye US Navy surveillance radar operating at ∼400 MHz;
  • US Air Force E-3 airborne warning aircraft command system (AWACS) employing a surveillance radar operating at ∼3 GHz;
  • Radar altimeters operating at ∼4 GHz, commonly used on civil and military aircraft;
  • Fighter aircraft operating in the 10–18 GHz range;
  • US Army AH-64 C/D Apache attack helicopter with Longbow radar (AH-64 D variant) operating at ∼35 GHz;
  • Active radar-guided, air-launched or ground-launched antiarmour missiles: either Hellfire or Brimstone operating at ∼94 GHz.

The entire frequency range used by radar and other radio applications is categorised by the letter identification scheme shown in Table 3.1. However, only those frequencies assigned by the International Telecommunications Union (ITU) are available for use. This categorisation does not mandate the use of a particular band or frequency but merely indicates that it is available to be used. Other factors decide which band to be used in a particular application: most notable are the effects of atmospheric absorption and the size of antenna that the platform can reasonably accommodate.

Figure 3.2 Range of electromagnetic applications in military avionics.

Figure 3.3 Airborne radar frequency coverage.

Table 3.1 Designation of radar bands [source: Skolnik, M.I. (1980) Introduction to Radar Systems, McGraw-Hill]

Band designatora Nominal frequency range ITU assignments
HF 3–30 MHz
VHF 30–300 MHz 138–144 MHz
216–225 MHz
UHF 300–1000 MHz 420–450 MHz
850–942 MHz
L 1–2 GHz 1215–1400 MHz
S 2–4 GHz 2300–2500 MHz
2700–3700 MHz
C 4–8 GHz 5250–5925 MHz
X 8–12 GHz 8500–10 680 MHz
Ku 12–18 GHz 13.4–14.0 GHz
15.7–17.7 GHz
K 18–27 GHz 24.05–24.25 GHz
Ka 27–40 GHz 33.4–36 GHz
V 40–75 GHz 59–64 GHz
W 75–110 GHz 76–81 GHz
92–100 GHz
mm 110–300 GHz 126–142 GHz
144–149 GHz
231–235 GHz
238–248 GHz

a IEEE Std 521–1984.

The effect of atmospheric absorption is a constraint depending upon physics that is totally outside the control of the designer. Physical antenna size is to some extent under the control of the designer, although the platform dimensions will be determined by factors relating to its airborne performance, range and so on. As for many systems, the design of a radar system is subject to many considerations and trade-offs as the designer attempts to reconcile all the relevant drivers to obtain an optimum solution.

The effects of atmospheric absorption are shown in Figure 3.4. The diagram illustrates the loss in dB per kilometre across the frequency spectrum from 1 to 300 GHz. This curve varies at various altitudes – the particular characteristic shown is for sea level. A 10 dB loss is equivalent to a tenfold loss of signal, so the loss per kilometre at 60 GHz is almost a 1000 times worse than the loss at around 80 GHz. These peaks of atmospheric absorption occur at the resonant frequency of various molecules in the atmosphere: H2O at 22 and 185 GHz and O2 at 60 and 120 GHz, with the resonance at 60 GHz being particularly severe.

Also shown on the diagram are four key frequency bands used by some of the weapons systems of today:

Figure 3.4 Effects of atmospheric attenuation.

  • Surveillance radar operating at ~3 GHz;
  • Fighter radar radiating from 10 to 18 GHz;
  • Attack helicopter operating at 35 GHz;
  • Anti-armour missile transmitting at 94 GHz.

It can be seen that the atmospheric absorption effects have a significant impact upon the portions of the spectrum that the radar designer can reasonably utilise.

The basic principle used by radar is portrayed in Figure 3.5. The energy radiating from a radar transmitter propagates in a similar fashion to the way ripples spread from an object dropped in water. If the radiated energy strikes an object – such as an aircraft – a small proportion of that energy is reflected back towards the radar. The transmitted energy effectively has a double journey: out to the target and back again. Radar uses this principle to measure the distance to the target; knowing that the speed of light is ≈3 ×108 m/s, and by measuring the time taken for the reflection to arrive, makes it possible to calculate the target range:

Figure 3.5 Basic principles of radar.

Figure 3.6 Pulse and continuous wave.\

where R is the range of the target, c is the speed of light (3 × 108m/s) and Δt is the time taken for the radar energy to perform the round trip.

Radar energy may also be transmitted in a number of ways. Figure 3.6 shows two situations; one where the RF energy is sent in pulses and the other where RF energy is radiated continuously – also known as a continuous wave.

Pulsed radar transmission is useful when information is required regarding the range of a target. Clearly, by transmitting a pulse of radar energy it is easy to measure when the reflected pulse returns and hence determine the target range using the formula given above.

Using a continuous wave transmission allows the closing (or receding) velocity of the target to be determined. This is achieved by using the Doppler effect. The Doppler effect is one by which the frequency of radiation is affected if a target is moving in the radial direction between radar and target (see Figure 3.7 which depicts a point radiating source travelling with a velocity from left to right).

If a radiating (or reflecting) target is receding from an observer, the frequency will appear to reduce as far as the observer is concerned. Conversely, if the target is approaching then the frequency will appear to increase. The classic illustration is of a train approaching, passing and receding from a stationary observer: as the train approaches, the sound pitch will be higher than when it recedes. As will be seen, the Doppler effect is a very useful property that is extensively used in various radar applications.

3.2 Radar Antenna Characteristics

In Figure 3.5 it is implied that the radar energy is directed in some manner in the direction of the target, and this is in fact the case. In order to achieve such directed energy, early systems used parabolic reflectors in which radiated energy is directed towards the reflector from a radiating horn at the focal point. Reflected energy returning from the target is ‘gathered’ by the reflector and concentrated at the focal point where the horn feed is situated. In this way the radiated energy is directed towards the target and the reflected energy is gathered from it, as illustrated in Figure 3.8.

Figure 3.7 Doppler effect due to motion.

In recent years, radars have adopted the planar array shown in the lower part of Figure 3.8. Whereas the parabolic reflector achieves beam shaping by means of its physical parabolic shape, the planar array achieves a similar effect by careful phasing within the RF feeds at the rear of the planar array. This principle is described in more detail in Chapter 4.

This radar antenna directional property is extremely important to the radar as it focuses the energy into a beam on transmission and effectively ‘gathers’ the reflected energy during reception. This directional property enhances the operation of the radar and is known as the antenna ‘gain’. The gain depends upon the size of the antenna and the frequency of the radiated energy.

The beamwidth of an antenna is actually quite a complex function, for, during the formation of the main beam, which is at the heart of the performance of the radar (owing to the antenna gain), a number of sublobes are also created which are distinctly unhelpful. These sublobes are known as sidelobes. Whereas the main beam is central to the performance of the radar, the sidelobes detract from the radar performance since they effectively waste energy and have other adverse effects.

Figure 3.8 Parabolic reflector and planar array.

Figure 3.9 Antenna pattern showing main beam and sidelobes.

As shown in Figure 3.9, the upper part of the diagram portrays the main beam and sidelobes generated by a typical directional antenna. The area of interest to the radar is the main beam. The sidelobes are characterised as the first sidelobe, the second sidelobe, etc., and for the antenna pattern shown there are five sidelobes, each becoming progressively weaker the further off-boresight (the radar centre-line) they are. Not only do the sidelobes waste energy during transmission by directing energy away from the target, they also allow stray and unwanted energy to enter the antenna and therefore the radar receiver during reception. Stray energy may be noise or ‘clutter’ produced by spurious reflections from the ground, alternatively it may be energy being transmitted by an enemy jammer who is attempting to confuse the radar. The actual beam pattern is determined by a specific mathematical relationship known as a sin x/x waveform. The beamwidth of the main beam is defined as the point at which the signal strength of the sin x/x waveform has dropped to 3 dB below (–3 dB) the peak value. In numerical terms this relates to a signal level 1/√2 below the peak signal which equates to 0.707 of the peak level.

The beamwidth varies according to the mode in which the radar is operating and theinformation it is trying to gather. The beamwidth also does not necessarily have to be the same in both axes (azimuth and elevation), as will be described. For an air-to-air mode the beamwidth will be narrow and be equal in azimuth and elevation, whereas for a ground-mapping mode it will be narrow in azimuth and broad in elevation.

The term decibel relates to a measurement unit that is used extensively within the radar community to describe relative signal levels in a short-hand logarithmic form according to a base of 10. In radar calculations, dynamic range between two signals may be several orders of magnitude, and continuously using many noughts is tiresome and confusing. The principle is shown in Figure 3.10.

Figure 3.10 Decibels compared with numerical format.

It can be seen that, as the signal strength increases to the right, so the decibels increase. At a positive signal ratio of 103 or 1000, the decibel figure is 30 dB [log10(1000) is 30], whereas for a decreasing signal ratio, a negative ratio of 10–3 or 1/1000, the decibel figure is –30 dB [log10(1/1000) is –30]. It is therefore easier to say that in a good antenna design the first sidelobe is 30 dB down (or –30 dB) on the main lobe signal than saying that the signal ratio is 1/1000th or 10–3 below the main lobe (this is a realistic figure in practice). Therefore, within radar terminology the decibel notation is used liberally to describe gains or losses when considering system performance.

3.3 Major Radar Modes

Examining the operation of some basic radar modes helps to understand how beamwidth and other factors such as pulse width, scan patterns, dwell time and pulse repetition frequency (PRF) are important to radar operation. The main modes that are described are:

  • Air-to-air search;
  • Air-to-air tracking;
  • Air-to-air track-while-scan (TWS);
  • Ground mapping.

3.3.1 Air-to-Air Search

One of the functions of a fighter aircraft is to be able to search large volumes of air space to detect targets. Many scan patterns are able to accomplish this function, but perhaps the most common is the four-bar scan shown in Figure 3.11. This scan comprises four bars stacked in elevation, and the radar mechanically scans from side to side in azimuth while following the four-bar pattern. The pattern shown begins in the top left-hand corner and finishes in the bottom left-hand corner before recommencing another cycle. The scan might typically cover ±30° in azimuth centred about the aircraft centre-line and about 10–12° in elevation. Alternatively, sector scans may be used, say ±10° skewed left or right off the centre-line if that is where the targets are located. The beamwidth in the air-to-air search mode will probably be ∽3°, and the scan bars will usually to be positioned one beamwidth or 3 dB apart to ensure that no target falls between bars. The search pattern is organised such that a target may be illuminated several times during each pass, as indicated by the overlapping antenna coverage shown in the figure. This allows the target to be detected with certainty and avoids nuisance detections or false alarms.

Figure 3.11 Typical air-to-air search pattern.

More importantly, integration techniques (summing the return from several successive pulses) allow the signal return to be enhanced and therefore the ability to detect the target in noise or clutter to be significantly improved. For similar reasons, integration techniques can enhance the range at which the target is initially detected. In earlier-generation radars, all the targets detected would be shown on the radar display and could lead to a confusing picture, particularly when trying to separate friend from foe. More modern radars with digital processors are able to categorise multiple targets more easily, thereby simplifying the engagement procedure.

It should be recognised that, when the radar is operating in this air-to-air search mode, enemy targets fitted with a radar warning receiver (RWR) or other detection equipment will know that they are being illuminated or ‘painted’ by the searching radar. Furthermore, by categorising radar signal parameters such as radiated frequency, pulse width and PRF, the enemy target will be able to identify what type of radar and what aircraft type is being encountered. The RWR will also give a bearing to the illuminating radar, and the ‘blip/scan’ ratio will indicate whereabouts in the radar scan pattern it is located. It will also be obvious that the radar energy only has to travel a single path to reach the potential target, whereas the energy has to travel out and back to the radar to produce a return. This means that the target aircraft will be receiving a much stronger signal than the radar. Therefore, while air-to-air search is a useful mode, radar operators should also be aware that at the same time they are also giving potentially useful information to their adversary.

3.3.2 Air-to-Air Tracking

On occasions the radar may need to obtain more pertinent data regarding the target, perhaps in order to prepare to launch an air-to-air missile. To attain this more specific target information, the radar needs to ‘lock on’ to the target. When this occurs the scan pattern changes and the radar antenna tracks the target in azimuth and elevation. The target is also locked in terms of range using a range gate. The radar is now able to track the precise movements of the target. In some tracking modes the PRF may be switched to higher frequency to increase the target data update rate (Figure 3.12).

Figure 3.12 Air-to-air tracking.

The target dataset will include the following data:

  • Range;
  • Azimuth;
  • Azimuth rate;
  • Target identification;
  • Range rate;
  • Elevation;
  • Elevation rate;
  • Target classification.

The accompanying changes in the radar characteristics detected by the potential target following lock-on is a warning that the engagement is becoming more serious. At this point the target may attempt evasive tactics – deploy countermeasures or chaff or jam the target radar.

The means by which the radar achieves target angle tracking and target range is described later in the chapter.

3.3.3 Air-to-Air Track-While-Scan

The disadvantage of locking on to the target and thereby signifying engagement intentions has already been described, but the advent of digital signal/data processing has enabled an elegant solution to be developed. Track-while-scan automates the process of deciding which target to engage (Figure 3.13). As TWS is under way, the radar processor progressively builds up a history of the flight path of targets within the scan. If successive measurements disagree, then the track is rejected; if the data agree, then the track is maintained. Gates are initiated that assign angular information, range and range rate to each track and predict where the target will be at the time of the next observation. If the track is stable, then the forecast gates will become more accurate and statistical filters will establish that the predicted fit is good. Techniques are used to arbitrate when gates overlap or where more than one target appears in the same gate perimeter.

Figure 3.13 Air-to-air TWS.

The advantages of TWS are as follows:

  1. Accurate digitised tracking data are established on each track within the antenna scan pattern without alerting potential targets that they are being tracked.
  2. The automation process allows many targets to be tracked accurately and independently.
  3. A typical radar using TWS will be able to track 20 or more targets in three-dimensional space.

3.3.4 Ground Mapping

From the early days of radar it was known that the radar could be used to map the terrain ahead of the aircraft. Using the different reflective characteristics of land, water, buildings, etc., it was possible to paint a representative map of the terrain ahead of the aircraft where major features could be identified. With the application of digital processing and advanced signal processing techniques, the ability to resolve smaller features increased and high-resolution mapping became possible.

Figure 3.14 Ground mapping.

While using the ground-mapping mode, the antenna sweeps from side to side as shown in Figure 3.14. The area illuminated by the mapping beam equates to the dotted boundary shown in the figure.

Whereas the air-to-air modes use a narrow pencil beam, a fan beam is used for ground mapping. That is, a beam where one dimension is narrow –2 or 3° – while the other is relatively broad, say 10 to 15°. The figure shows that the ground-mapping beam is narrow in azimuth and wide in elevation; this represents the optimum shape for the mapping function.

This describes the operation of a basic ground-mapping mode. In the past 20 years, improved capability and flexibility have been achieved by the use of digital computing in the radar data processor (RDP) and presignal processor (PSP). By the use of fast Fourier transform (FFT) for signal processing and aircraft motion compensation, increasingly sophisticated radar modes have been developed. These include:

  • Doppler beam sharpening (DBS);
  • Synthetic aperture radar (SAR);
  • Inverse Synthetic Aperture Radar (ISAR).

These modes will be described in Chapter 4.

3.4 Antenna Directional Properties

Earlier it was stated that the directional properties of an antenna were determined by the radiated frequency and the size of the antenna. There are simple formulae that help to estimate the beamwidth and gain of an antenna if these parameters are known.

The frequency and wavelength of an electromagnetic wave are related to each other and the speed of light, c, by the equation

where c is the speed of light (3 × 108 m/s ), f is the frequency (Hz) and λ is the wavelength (m).

Therefore, for the airborne fighter radar described earlier, operating at a frequency of 10 GHz (1010 Hz), λ=c/f=3 × 108/1010 = 3/100 or 0.03 m or 3 cm.

Another formula is a ready approximation to determine the beamwidth θ of an antenna knowing the frequency and the antenna size:

where θ is the beamwidth (deg), λ is the wavelength (m) and D is the antenna dimension (m).

Using again the example of the airborne fighter radar, and assuming an antenna dimension of 0.6 m (~ 24 in), θ≈6 × 0.03/0.6 ≈ 3.5°.

Using a similar approximation, it is possible to estimate the antenna gain:

where GD is the antenna gain, θB is the beamwidth (rad) in one axis and φB is the beamwidth (rad) in the orthogonal axis (one radian ≈57.3°).

Again using the fighter radar example, GD = 4 × π × (57.3)2/(3.5)2 ≈ 3368. This gives an idea of the advantage that the antenna gain confers. Expressed in decibels, the antenna has a gain of log10(3368) or around 35 dB.

3.5 Pulsed Radar Architecture

The basic principles of radar operation have already been outlined. The detailed operation of pulse radar is described in this section. A top-level diagram of a pulsed radar system is shown in Figure 3.15.

3.5.1 Pulsed Radar Components

The diagram shows the major elements which are:

  • Modulator;
  • Transmitter;
  • Antenna;
  • Receiver;
  • Video processor.

Figure 3.15 Top-level pulsed radar architecture. Modulator

The modulator determines the pulse shape and the nature of the radar modulation. Although pulsed transmission is the most elementary form of radar operation, the modulation in a modern multimode radar may take many forms depending upon the nature of information being sought. The operation of the modulator is controlled by the synchroniser which dictates when a pulse should be initiated. The modulator uses the superheterodyne (‘super-het’) principle of modulation to superimpose the modulating signal upon the high-frequency carrier to provide a composite waveform. Transmitter

The transmitter amplifies the modulated carrier signal and feeds it to the antenna via a duplexer. This serves the function of directing the transmitter energy to the antenna waveguide system to be fed by the antenna elements for transmission into the atmosphere. It also routes the reflected target energy to the receiver. Antenna

The antenna, as has been described, directs the radar energy towards the target and receives the reflected energy from the target. Along with the target echo, a substantial amount of clutter from ground returns is also received. The antenna beam is focused according to the shape of the antenna and the nature of the beam required. Unwanted radar energy enters through the antenna sidelobes as well as the main beam. The antenna also receives noise from a variety of external sources that can help mask the true target signal.

Returning energy is passed through a receiver protective device which blocks the large amounts of transmitted power that would cause severe damage to the receiver, but also at the appropriate time allows the reflected target energy to pass through. Receiver

The receiver amplifies the reflected target signal and performs the demodulation process to extract the target data from the surrounding noise, and the resulting target video data are passed to the video processor. Video Processor

The video processor is also controlled by the synchroniser in order that transmitted pulse and target return pulses are coordinated and that a range measurement may be made. The resulting data are coordinated and displayed on the radar display.

3.5.2 Pulsed Modulation

The nature of the pulse modulation in terms of pulse width and frequency of repetition is highly interactive with a number of important radar characteristics and has a significant impact upon the performance of the radar. The basic parameters of a pulsed radar signal are described in Figure 3.16.

In pulsed radar operation, the carrier frequency is modulated by the envelope of a single rectangular pulse; in this case the pulse embraces a fixed carrier frequency. As will subsequently be discovered, in sophisticated radar operations there are advanced forms of modulation/transmission in which the pulse is not rectangular nor the carrier fixed in frequency. The pulse width is denoted by the symbol τ and is usually fairly narrow, perhaps ∼1 μs in an air-to-air mode. After a time interval called the pulse period, a second pulse is transmitted and the sequence is repeated. The rate at which the pulses are repeated is called the pulse repetition frequency (PRF), and both the pulse width τ and the PRF are key radar parameters.

Figure 3.16 Pulsed radar transmission.

It will be noted on the diagram that the terms ‘time domain’ and ‘frequency domain’ are mentioned. The time domain is familiar in everyday life as it is the domain in which we live; the frequency domain is more abstract but is of great importance to the radar designer. In fact, the time and frequency domains are interdependent and interwoven, and this has a significant impact upon the operation of radar systems. What happens in the time domain affects the frequency domain, and vice versa.

The rectangular pulse τ results in a response in the frequency domain that has a sin x/x response, the same generic response that determines the pattern of the antenna main beam and sidelobes. However, in this case the response is occurring on an axis relating to frequency rather than angle off-boresight as is the case in the antenna pattern. When the incoming pulse is received, it results in a frequency response of received power portrayed by the sin x/x response and centred upon the radiated frequency f0. The practical limits of the main sin x/x response are ±f1 centred on fo, that is, f0 + 1/τ; f0 – 1/τ (f0 being the carrier frequency and τ the pulse width), and this determines the bandwidth required of the receiver in order to be able to pass all the components of the target return. Therefore, for a 1μs pulse the receiver bandwidth would need to be 2/τ = 2/(1 × 10–6) = 2 × 106 or 2 MHz (see the lower part of Figure 3.16). The narrower the transmitted pulse, the wider will be the bandwidth; the converse also applies.

In basic pulsed radar operation the pulse width also determines the range resolution of which the radar is capable. The radar can only resolve to half the pulse width, as at a lower interval than this part of the pulse has been reflected while part has not yet reached the target. A pulse of 1 μs duration will be approximately 1000 ft long (as light travels at 3.3 × 3 × 108 or ∼109 ft/s and the duration of the pulse is 1 × 10–6; distance = velocity × time). Therefore, a 1μs pulse will be able to resolve the target range to no less than 500 ft. In fact, by using more complex modulation and demodulation methods called pulse compression, it is possible to achieve much better resolution than this; pulse compression is discussed in Chapter 4.

The pulse period – and therefore PRF – also has an impact upon the radar design that affects target ambiguity, as shown in Figure 3.17. The figure shows an aircraft illuminating two targets and compares the effect of the returns from these targets for two different pulse periods. In pulse period T1 (PRF1 = 1/T1), the returns from both targets are received before the successive pulse is transmitted, and the range is unambiguous. In the case of the shorter pulse period T2 (PRF2 = 1/T2), the return from the most distant target occurs after the transmission of the successive pulse and to the radar appears as a relatively close target within the second period. In this case the target range is ambiguous and misleading. The selection of PRF is one of the most difficult choices the radar designer has to make, and some of the effects of range ambiguity are discussed in more detail in Chapter 4.

The fact that the radar only transmits for a portion of the time means that the average power is relative low. The average power is given by the following expression:

where Ppeak is the peak power, τ is the pulse width and T is the pulse period.

Figure 3.17 Effect of pulse period on target ambiguity.

For a peak power of 10 kW, a pulse width of 1 μs and a pulse period of 250 μs, we have

3.5.3 Receiver Characteristics

In order to be able to detect the target, the radar receiver has to able to discriminate from unwanted effects. The main adverse affects are as follows:

  1. Noise is either internally generated or radar transmitter induced or externally sourced. Noise is random and can only be minimised by good design.
  2. Clutter due to unwanted returns from the ground and other sources is usually more systematic and can be countered by filtering and processing techniques. Noise

The sources of noise that can affect the ability of the radar receiver to detect a target signal are shown in Figure 3.18. The total system noise includes noise from the following sources:

  1. Antenna noise Ta. The antenna noise includes all those sources of noise that are external to the radar, including radiation from the sun, terrain, emissions from man-made objects and the weather. Noise from jamming may also be included in this category. The radome and the antenna itself may also generate noise. While external noise will be most troublesome when it enters the system via the antenna main beam, it should also be remembered that noise can also enter via the antenna sidelobes.
  2. Transmission line noise Tr. This includes noise originating within the waveguide couplers, duplexer and the receiver protection device.
  3. Equipment noise Te. The equipment noise is generated within the receiver itself and is the most difficult to counter.

Figure 3.18 Sources of noise affecting radar signal.

The total system noise, Ts, is the sum of these individual components:

The problem with the noise in a receiver is that, once present, it is there to stay. Signal amplification in subsequent stages will only amplify the noise as well as the signal and accentuate the problem of target detection. One technique commonly used is to insert a low-noise amplifier (LNA) at the front of the receiver to amplify the signal proportionately more than the noise. LNAs are also commonly used where antennas (or apertures) are mounted remotely throughout the airframe and where transmission losses might be relatively high.

The receiver noise is defined as noise per unit of receiver gain:

The receiver gain can be easily measured using laboratory techniques.

The receiver noise may be characterised by a figure of merit or noise figure Fn. This is defined as the ratio of the noise figure of the actual (imperfect) receiver to the hypothetical ideal receiver providing equal gain. Therefore:

An ideal receiver would produce no noise; the only noise that would exist would be that from external sources. This external noise can be represented as though resulting from thermal agitation in a conductor (resistor) since the two have similar spectral characteristics. Therefore, in the derivation of Fn, for both ideal and actual receivers, the thermal noise can be portrayed as the voltage across a resistor. Thermal noise is governed by the random motion of the free electrons within the conductor and is uniformly spread across the entire spectrum. This motion is determined by the absolute temperature of the notional resistor, denoted by T0. Also, the noise depends upon the receiver bandwidth B. Thus, to derive the mean noise power for an ideal receiver, the expression

may be used for an ideal receiver, where k is Boltzmann’s constant = 1.38 × 10–23 W s/K, T0 is the absolute temperature of the resistor representing the external noise (K) and B is the receiver bandwidth (Hz).

The external noise is the same for both receivers, and by convention T0 is taken to be 290 K which is close to room temperature. Where the external noise is small by comparison with that generated by the receiver, as is usually the case, the mean noise figure for an actual receiver may be determined by the following:

As was shown earlier, the total noise may represented by Ts where the mean noise power (all sources) = k × Ts ×B.

The nature of the modulation used also has an impact upon receiver noise. This is shown in Figure 3.19. The figure shows the simple comparison of narrow and broad rectangular pulse modulation. It was shown earlier in Figure 3.16 that the bandwidth needed to accommodate all the frequency components of a rectangular pulse was governed by the sin x/x waveform, and that the theoretical bandwidth was 2/τ. The narrow (sharper) pulse τ1 needs a greater bandwidth than the broader pulse τ2. The narrow pulse gives an improved range resolution and, for a given pulse period (PRF), a reduced mean power, so it can be seen that there are performance trade-offs to consider that affect bandwidth and hence receiver noise.

In practical systems a compromise is allowed and generally a bandwidth of 1/τ is regarded as sufficient. Therefore, it is common practice to narrow the IF bandpass filter until it is 1/τ wide, just wide enough to pass the bulk of the target-related energy but reject the unwanted noise. This design is called a matched filter, and the mean noise energy per pulse is kT0/τ.

In the Doppler radars addressed in Chapter 4 the Doppler filters downstream of the IF filter are much finer, and greater noise and clutter rejection result.

Figure 3.19 Effect of different pulses on the receiver bandwidth.

The detection and extraction of a target echo from a background of noise depends upon the four factors outlined below:

  • The average power radiated in the direction of the target;
  • The proportion of the radiated energy reflected back in the direction of the radar;
  • The proportion of power recaptured by the radar antenna;
  • The length of time the antenna beam is trained upon the target.

Average power is determined by the relationship of the peak power, Ppeak, transmitted by the radar and the modulation characteristics of pulse width, τ, and pulse period, T, as shown in the previous section. The antenna gain, GD, also increases the power density related to the beamwidth(s) and beam geometry.

As the radiated signal is directed towards the target, it spreads out an increasing area, proportional to R2, where R is the range from the radar. This means that the power density reduces by a factor of 1/R2 as the energy is propagated in the direction of the target.

A fraction of the energy incident upon the target will be reflected back in the direction of the radar. In the simplest form the target may be considered to be a simple sphere with a specific cross-sectional area, denoted by the symbol σ and specified in square metres. The reality is much more complicated than that, and other factors such as reflectivity and directivity play a great part, as will be seen in the discussion on low observability or stealth in Chapter 4.

As the energy is reflected back to the target, the 1/R2 effect applies in terms of the reduction in received power density. The impact of this effect means that the energy received at the radar has been reduced by a total factor of 1/R4 in its outward and return path to and from the target. This has an impact upon the ability of the target signal to be detected above the noise, as shown in Figure 3.20. The figure shows how the returning signal (not to scale) decreases with increasing range to the point where the signal is not detectable against the noise background.

Figure 3.20 Effect of range upon the target echo.

As will be seen, the equation governing the strength of the return signal is a fourth-power law, and this means that the receiver has to accommodate a very large dynamic excursion in terms of variation in target signal strength as the range varies. In certain modes this is addressed by a technique called sensitivity time control (STC) in which the receiver gain is reduced at very short ranges and increased progressively during the range sweep. This technique is sometimes referred to as swept gain and to some extent mitigates the problem of extremely high signal returns at short range.

Another technique is often used to counter this effect and prevent the receiver amplifiers from saturating: if the receivers saturate, then both signal and noise will merge as the amplifiers clip both noise and target signal returns. In this case, automatic gain control (AGC), as the name suggests, automatically reduces amplifier gain to prevent saturation occurring.

The actual detection of the target signal is determined by the setting of a target detection threshold as shown in Figure 3.21. This shows two targets, A and B, against a background of noise on a time axis: A and B are obviously at different ranges from the radar. The figure shows the importance of setting the target threshold correctly with respect to the mean noise level. If the threshold is set low, then it may be anticipated that more targets may be detected. However, as the diagram shows, setting a low target threshold has the accompanying risk of detecting a spurious target – called a false alarm. For the low threshold setting shown, the radar would detect three targets: genuine targets A and B and the false alarm.

Conversely, there are problems with setting the threshold too high to avoid false alarms. In this case the return from genuine target A is lost and only target B is detected.

One of the major factors affecting target detection was antenna time on the target. So far, only the detection of a target using a single pulse has been considered. In fact, as the radar beam sweeps through the target, a number of successive pulses will illuminate the target in a short period of time. Most radars have the capability of integrating the detected output over a number of pulses, and this has significant advantages, as can be seen from Figure 3.22.

Figure 3.21 Receiver threshold setting.

Noise is generally random in terms of amplitude and phase. The target return is more systematic and repetitive in nature, at least over a range of successive pulses in an antenna scan. The effect of integrating noise over a series of pulses is to end up with noise at more or less the mean noise level before integration. The converse is true for a real target return. The target return is aggregated during the integration process and the result is a much stronger target return. The figure shows that as an example – integration over 12 pulses produces an integrated signal that comfortably exceeds the target detection threshold, whereas the integrated noise does not. This occurs in spite of the fact that each of the individual target signals are well below the target detection threshold and without pulse integration would each be subsumed by noise. This shows the powerful capability of ‘extracting’ a signal from noise using integration techniques.

Figure 3.22 Effect of integration over several pulses.

The actual antenna time on target depends upon a combination of three factors:

  • The antenna scan or slew rate;
  • The antenna beamwidth;
  • The PRF.

Taking some simple figures by way of illustration, if the antenna scan rate is 60 deg/s and the 3 dB beamwidth is 3°, then the antenna will dwell upon a target for 1/20th of a second. If a medium PRF of 1000 Hz is assumed, then the antenna will theoretically have a total of 50 ‘hits’ on the target during every pass across the target. Clutter

The effects of clutter, particularly from ground returns or precipitation, can cause large amounts of unwanted signal being returned to the receiver. Clutter can enter the receiver channel through the main beam or via the sidelobes. It can depend upon the nature of the terrain, terrain geometry and the aspect (depression angle) of the antenna boresight. If the clutter is from the water, then it may depend upon sea state (the height of the waves or the smoothness of the water surface). In some ways, clutter may be systematic in terms of the effect that it has on the radar, and in these cases it is easier to counter or filter.

Moving targets, or targets with a significant radial velocity with respect to the radar, may have a Doppler shift component that may enable the target to be distinguished against a stationery background. The use of Doppler filters and target velocity techniques is described in Chapter 4.

3.5.4 Radar Range Equation

The foregoing discussion leads us to the equation that is the most powerful and commonly used when examining the performance of radar systems, that is, the radar range equation. The radar range equation takes many forms depending upon those factors that need to be taken into account and the type of transmission being considered. In the simplest form, the maximum range for a single radar pulse is determined by the following equation:

where R is the radar range (m), Ppeak is the peak power (W), G is the antenna gain (m2) (this may also be expressed in decibels for ease of calculation, as explained earlier), σ is the target cross-sectional area (m2), τ is the transmitted pulse width (s), and Smin is the minimum detectable signal energy (W-s). This equation does not take account of pulse integration.

There are some interesting observations to make regarding this formula:

  1. Peak power Ppeak. As the peak power only affects the radar range by the inverse fourth power, doubling the peak power of the radar only increases the range by the inverse fourth power of 2 ≈1.19 or 19%.
  2. Antenna gain G. If the antenna is circular, doubling the size of the antenna will increase the gain of the antenna by 4, and the overall range by a factor 1/2 or by about 71%. However, commensurate with the antenna gain, the beamwidth would halve, which may make target acquisition more difficult. Dwell time might also have to increase to improve target integration. Altering the wavelength of the radiated transmission would have an effect upon radar range as the range alters by the inverse square of the wavelength. Decreasing wavelength or increasing frequency can therefore increase the range. The atmospheric absorption outlined in Figure 3.4 earlier in the chapter will be an important factor, as at certain parts of the spectrum absorption rates are punitive, more than cancelling out any benefit that increasing radiated frequency may confer.
  3. Target cross-sectional area σ. Reducing the target cross-sectional area by a factor of 60 dB (equivalent to 1 × 10–6) by using extensive low observability (LO) techniques reduces the range by a factor of ∽30.
  4. Pulse width τ. Maintaining mean power but decreasing the pulse width increases the peak power but also increases the receiver bandwidth, allowing more noise into the receiver.
  5. Minimum detectable signal Smin. Decreasing the minimum detectable signal increases the radar range, but the risk of false alarms may increase.

As more factors are taken into account, so more trade-offs need to be made. However, as will be seen later, the adoption of sophisticated modulation and signal processing techniques can gain significant performance enhancements in modern digital radars.

3.6 Doppler Radar

In the early part of the chapter the Doppler effect was described, that is, the effect upon radiated frequency when a moving source approaches or recedes from an observer. The same effect occurs when radar energy is reflected by ground clutter, except that the Doppler frequency shift is doubled as the radio energy has to travel out and back to the radar. Normally, ground returns are a nuisance as far as the radar is concerned, and all means are used to reject the ground clutter. However, there is one radar application where the ground clutter Doppler frequency shift is utilised, and that is the Doppler radar, sometimes called the Doppler navigator. A typical configuration for a Doppler radar is shown in Figure 3.23.

The Doppler radar comprises three or four narrow, continuous wave radar beams angled down from the horizontal and skewed to the left and right of the centre-line. The three-beam layout shown in the figure is called a lambda configuration for obvious reasons. The diagram shows a situation where the aircraft is flying straight ahead, i.e. heading equals track and there is no angle of drift because of crosswind. The forward beams 2 and 3 will experience a positive Doppler shift as the ground is advancing towards the aircraft. The Doppler shift Δf is proportional to 2V/ λ, where V is the aircraft forward velocity and λ is the wavelength of the radiated frequency. The aft beam 1 will experience a negative Doppler shift proportional to 2V/λ as the ground is receding from the aircraft. There are several scaling factors including direction cosines associated with the beam, but subtracting forward and aft beams yields a signal proportional to 4V/λ. Therefore, by manipulating and scaling the Doppler shifted returns from all three beams, the aircraft horizontal velocity with respect to the ground (i.e. ground speed), Vx, may be calculated.

Figure 3.23 Doppler radar.

If the aircraft is drifting left or right owing to a cross-wind, then, by using the cross-track Doppler shift components and a similar manipulation process, the cross-track velocity, Vy, may be calculated. The vertical velocity component, Vz, may also be calculated. The vector sum of Vx, Vy and Vz enables the total aircraft velocity, V , to be established. Doppler radars do have one disadvantage: if the terrain is very flat with a low reflectivity coefficient, then insufficient energy may be reflected back to the radar and the Doppler shift cannot be measured. Such effects can be achieved when travelling over very calm water or ice-covered expanses of water. Doppler radars were very commonly used before inertial navigation systems (INS) became the norm about 30 years ago; more recently, INS has been augmented by the satellite-based global positioning system (GPS). The initial avionics configuration of Tornado included a Doppler radar, and they are still frequently used on helicopters as air data becomes very unreliable at low airspeeds. For further data on Doppler radars, see Kayton and Freid (1997).

As will be explained in Chapter 4, sophisticated radars in use today combine the use of pulse techniques and Doppler to produce pulsed Doppler (PD) modes of operation.

3.7 Other Uses of Radar

3.7.1 Frequency Modulation Ranging

The use of pulsed radar techniques has hitherto been described to measure target range. However, frequency modulation may also be used to determine range as depicted in Figure 3.24.

Figure 3.24 Frequency modulation ranging.

The transmitted signal consists of a triangular wave modulation, as shown, that sweeps across the frequency spectrum, completing one cycle in 0.01 s in the example given. The received frequency will lag the transmitted frequency by an amount Δf owing to the time taken to complete the out and return journey. The example shows a measurement taken when the reflected received frequency, f1, is compared with the current frequency at the transmitter, f2, with the difference in frequency being Δf . The associated time difference signal, Δt, is proportional to the range of the target.

The figures shown on the diagram relate to the use of this technique in a radar altimeter, where the radar returns are used to calculate the instantaneous altitude of the aircraft above the terrain over which the aircraft is flying. In this example, the transmitter is sweeping in a linear manner over a frequency range of 4250–4350 MHz in 0.01 s. The use of radar altimeters is described in Chapter 7.

3.7.2 Terrain-following Radar

Whereas the radar altimeter is useful in informing pilots where they are in relation to the terrain underneath the aircraft, it does not tell them where the terrain is in front of the aircraft. To do this, the pilot needs to use a terrain avoidance (TA) mode or, better still, a dedicated terrain-following radar (TFR). The TA function can be crudely achieved by using a normal pulsed radar in a single-bar scan mode with a fixed depression angle. This will tell the pilot where he is in relation to the terrain ahead of the aircraft, but it is not a sophisticated mode and does not readily lend itself to coupling into the autopilot (Figure 3.25).

The TFR is a dedicated radar coupling into a dedicated functional system and autopilot that allows the pilot much greater performance and flexibility when penetrating at low level at night. The TFR scans the terrain ahead of the aircraft and receives ground returns that are used for guidance. Normally, a simple box scan is used where the active sweeps are those in the vertical direction (sections 1 and 3). In some circumstances a figure-of-eight scan is used which provides broader lateral coverage than the simple box scan. The TFR therefore builds up a range/elevation picture of the terrain ahead of the aircraft and calculates an imaginary ‘ski-toe’ profile that reaches out ahead of the aircraft. This profile is calculated taking into account such factors as aircraft speed, manoeuvrability, etc., and provides an envelope within which the aircraft will not be able to avoid the terrain ahead. The system is configured so that, whenever the terrain ahead broaches the ski-toe envelope, the aircraft pitches up to rectify the situation. Similarly, if the terrain drops away in front of the aircraft, the aircraft pitches down until just operating outside the profile. The system operates just like the toe of a ski, moving up or down to follow the terrain ahead of the aircraft but always ensuring the aircraft can safely manoeuvre.

Figure 3.25 Terrain-following radar operation.

The measurements from the radar altimeter are also fed into the terrain-following system which calculates the ‘most nose-up command’ provided by either TFR or radar altimeter. This has the advantage of providing the pilot with an additional altitude safety buffer directly beneath the aircraft as the TFR is looking several miles ahead.

The TFR/radar altimeter commands may be coupled into the autopilot to provide an auto-TF mode while the aircraft is approaching the target area, thereby enabling the aircraft to fly at low level automatically while the pilot performs other mission-related tasks. The TFR may be an embedded system forming part of the aircraft primary radar, alternatively it may be provided in a pod that is loaded on to the aircraft. The AN/AAQ-13 LANTIRN navigation pod fitted to F-15 and F-16 aircraft performs a TFR function for these aircraft.

3.7.3 Continuous Wave Illumination

On some weapons systems a continuous wave (CW) illumination mode is provided. This mode is used when aircraft are fitted with semi-active air-to-air missiles; that is, missiles that can receive incoming RF energy and once fired can track and engage the target. As the missiles are unable to transmit, the aircraft radar has to provide the target illumination and it does this by using a CW illuminator co-boresighted with the aircraft radar antenna. Therefore, when the aircraft radar is locked on to the aircraft it can simultaneously illuminate the target (Figure 3.26). The disadvantage of this technique is that the aircraft radar has to remain locked on to the target and transmitting CW illumination until the engagement is complete. In high-density air-to-air combat this may not always be possible.

Figure 3.26 CW illumination.

3.7.4 Multimode Operation

Modern radars such as those on the F-15E and F-22 have the capability of operating simultaneously in a number of modes, an example of which is shown in Figure 3.27. In this hypothetical example, three simultaneous modes are depicted:

Figure 3.27 Simultaneous multimode operation.

  • Sector ground mapping;
  • Synthetic aperture (SA) spot mode;
  • Track-while-scan (TWS) mode engaging three separate targets.

The radar achieves this capability by interleafing the radar modulation required for each mode on a pulse-by-pulse basis and effectively operating as several radars in one. This offers immense flexibility to the aircraft as a weapons platform.

3.8 Target Tracking

During the pulsed radar tracking mode when the radar is locked on, it follows and automatically maintains key data with respect to the target:

  • Tracking in range;
  • Angle tracking in azimuth and elevation.

Tracking is maintained and the radar is said to have ‘target lock’ when all these loops are closed.

3.8.1 Range Tracking

Tracking in range is usually accomplished using a technique called range gating which automatically tracks the target as its range increases or decreases. The concept of the range gate is shown in Figure 3.28.

The radar return in the region of the target return will comprise noise and the target return. The range gating technique uses two gates, an ‘early gate’ and a ‘late gate’. The early gate is positioned near the leading edge of the target echo and detects and captures energy from the early part of the target return. Conversely, the late gate is positioned near the trailing edge of the target echo and detects and captures the energy from the trailing edge of the target return.

Figure 3.28 Range gate tracking.

The detected signals from the early and late gate are compared and the result is used to position the tracking gate so that it is coincident with the target return. In the example shown, both the early and late gates are positioned early (to the left) of the target return and the tracking gate is also incorrectly positioned. Consequently, the early gate detects less energy than the late gate. Identifying this discrepancy will cause the energy from early and late gates to be equalised and the tracking gate to be moved to the right (down range) so that it correctly coincides with the target echo. While the radar maintains target lock this process will be continued, maintaining the tracking gate at the same range as the target echo.

3.8.2 Angle Tracking

During the radar tracking mode the radar tracks the angle to the target in azimuth and elevation. In other words, the line-of-sight (LOS) to the target and the radar boresight are kept as close as possible. The LOS needs to be established within a frame of reference and usually the radar is stabilised in roll and pitch using attitude data from the aircraft attitude sources: inertial reference system (IRS) or secondary attitude and heading reference system (SAHRS). The final axis in the orthogonal reference set is usually the aircraft centre-line/heading.

There are three main methods of angle tracking that are commonly used, these are:

  • Sequential lobing;
  • Conical scan (conscan);
  • monopulse. Sequential Lobing

One of the first tracking radar principles adopted was sequential lobing, which in its earliest form was used in a US Army angle tracking radar air defence radar. The principle of operation of sequential lobing is shown in Figures 3.29a and 3.29b.

To track a target in one axis, two lobes are required; each lobe squints off the radar boresight. The centre point of where the two lobes overlap represents the boresight of the antenna and this is the LOS that the radar antenna is trying to maintain. It can be seen that, when the signal return from the target is the same in both beams, the LOS to the target has been achieved. As the target moves, continual error signals will be sensed and the antenna servo system responds by nulling the error and maintaining LOS to the target. If four lobes, A and B and C and D, are positioned as shown in Figure 3.29a, then lobes A and B provide tracking in elevation and lobes C and D provide tracking in azimuth. The reflected signal from the target received in each of the four lobes is routed via a channel switching assembly sequentially switched into the receiver. In this way, each of the four lobe returns is measured and error signals are derived to drive the antenna elevation and azimuth drive servomotors.

Figure 3.30 illustrates how each of the four lobes is switched in turn into the receiver. In practice, the waveguide switching arrangement is cumbersome and prone to losses; therefore, radar performance is compromised. The other significant disadvantage suffered by this method relates to the time taken for the sequencing to occur. The radar PRF will determine the maximum time that the receiver will be switched to a particular lobe. Only when the radar has completed the range sweep for a particular PRF can the receiver sequence to the next lobe. Furthermore, the elevation and azimuth error can only be updated once per cycle, and this adversely affects update rate and tracking error.

Figure 3.29 Sequential lobing – principle of operation.

Sequential lobing can be detected by a target and transmissions can be devised that will cause the radar to break lock. Transmitting on all four beams and receiving only on one may counter this. This technique is called lobe on receive only (LORO). Conical Scan

Conical scan (conscan) is the logical development of the sequential lobbing scheme already described. Conscan is another example of the earliest form of angle tracking – used because it was the easiest to use with the technology available at the time. The concept of conscan is depicted in Figure 3.31.

Figure 3.30 Sequential lobing – tracking configuration.

In conscan, one lobe is used that squints off-boresight. The lobe is then rotated such that the target return is enclosed within the imaginary cone that is swept out around the antenna boresight. In a parabolic antenna the rotating conscan beam is achieved by rotating the antenna feed at the desired rate. In more sophisticated arrangements the feed may be nutated, and this can achieve a better tracking performance than the straightforward rotating feed at the expense of a more complex feed mechanism. Typical conscan rates may be in the range up to 50 Hz.

Figure 3.31 Conical scan – principle of operation.

Figure 3.32 Conical scan – tracking configuration.

If the target is located off-boresight, the receiver will receive an amplitude-modulated signal at the conscan frequency. By detecting the signal and resolving the error components, drive signals are fed to the azimuth and elevation servo motors to move the antenna such that the target is back on boresight. The architecture of a conscan tracker is shown in Figure 3.32. As for the sequential lobing technique, there is only one receiver, but this is continuously fed with the conscan signal and the sequencing delays experienced with sequential lobing are avoided.

Conscan does offer these performance advantages over the sequential lobe technique but does itself suffer from a major disadvantage. Potential foes can identify the conscan frequency and can radiate a signal modulated with the conscan frequency that can cause the radar to break lock. Conscan is therefore susceptible to electronic countermeasures. This deficiency may be overcome if the target is illuminated with a non-scanning beam and conscan is used only for the receive channel. In this way, adversaries do not know they are being tracked by conscan means since the tracking operation is opaque to them. This technique is known as conical scan on receive only (COSRO). Monopulse

Monopulse is the preferred tracking method, and most tracking modern radars use it out of choice. The term monopulse means that a tracking solution may be determined on the basis of a single pulse rather than the beam sequence (sequential lobing) or a complete conical scan. The tracking data rate is therefore much higher and therefore potentially more accurate. Another advantage is that the tracking is based upon the simultaneous reception of the target return in all four channels and any variation in the echo in time can be readily accommodated. This is not the case with the other techniques.

Monopulse uses four simultaneous beams as shown in Figure 3.33, in which the beams are stacked in elevation and side by side. All four beams squint away from the antenna boresight by a small amount. Comparison of the target returns in all four channels is undertaken and error signals are derived to drive the antenna azimuth and elevation drive servo motors as appropriate. Monopulse techniques may use either phase or amplitude comparison to perform the tracking task. Of the two, amplitude comparison is generally preferred. A former UK AI radar (AI23b) used in the Lightning aircraft employed amplitude comparison in the elevation channel and phase comparison in the azimuth channel.

Figure 3.33 Monopulse – principle of operation.

All four channels transmit the same signal. The target return is received in each part of the monopulse array and fed into waveform junctions called hybrids which perform the sum and differencing function. A simplified portrayal of this arrangement is shown in Figure 3.34. Downstream of the summing and differencing process, three RF channels are formed: the sum channel (A + B + C + D); the elevation difference channel (A + B) – (C + D) ; the azimuth difference channel (A + C) – (B + D) . Each channel is fed into the receiver which has three corresponding channels. The sum channel is used to measure range, and the elevation and azimuth difference channels are used to drive the antenna elevation and azimuth servo drives respectively.

Figure 3.34 Monopulse – tracking configuration.

In the early days of monopulse radar the provision of three identical receiver channels caused problems and some compromises were sought that multiplexed two channels. The radars of today do not experience this problem.


Kayton, M. and Freid, W.R. (1997) Avionics Navigation Systems, 2nd edn, Wiley-Interscience.

Lovell, B. (1992) Echoes of War – The Story of H2S Radar, Adam Hilger.

Pilot’s Handbook – Honeywell Radar RDR-4B.